1. Technical Field
The present invention relates to class-G amplifiers. More particularly, the present invention relates to a compensation technique for a class-G amplifier that stabilizes frequency responses and greatly reduces the transient created by traversing the switching points between a low power supply and a high power supply.
2. Related Art
Class-G amplifiers operate to change the power supply voltage from a lower level to a higher level when larger output swings are required. Class-G operation is often implemented with a single class-AB output stage that is connected to two power supply rails by a diode, or a transistor switch. The design is such that the output stage is connected to the lower supply voltage, and automatically switches to the higher power supply rails for large signal peaks. Another approach involves the use of two class-AB output stages, each connected to a different power supply voltage, with the magnitude of the input signal determining the signal path. Using two power supplies improves efficiency enough to allow significantly more power for a given size and weight.
FIG. 1 shows a schematic diagram of a typical class-G current feedback amplifier. The input stage of the amplifier of FIG. 1 includes transistors 2, 4, 6, and 8, and current sinks 10 and 12. The collector terminals of transistors 2 and 4 are connected directly to an upper high voltage power supply terminal Vsph and a lower high voltage power supply terminal Vsmh, respectively. The emitter terminal of transistor 2 is connected through a current sink 12 to Vsmh, while the emitter terminal of transistor 4 is connected through a current sink 10 to Vsph. The base terminals of transistors 2 and 4 are connected together and form the non-inverting input (+input) of the class-G amplifier. Transistor 6 has a collector terminal connected to the input of a pull-up current mirror 18, an emitter forming the inverting input (xe2x88x92input) of the class-G amplifier, and a base terminal connected to the emitter terminal of transistor 4. Transistor 8 is connected in an emitter follower configuration with transistor 6, with a collector terminal connected to pull-down current mirror 20, and a base terminal connected to the emitter terminal of transistor 2.
The output stage of the class-G amplifier of FIG. 1 includes current mirrors 18 and 20, transistors 30, 32, 34, and 36, diodes 14, 16, 38 and 40, and voltage supplies 26 and 28. The collector of transistor 30 is connected to Vsph, while the collector of transistor 36 is connected to Vsmh. The emitters of transistors 30 and 36 are connected to the collector terminals of common emitter transistors 32 and 34, respectively. The base of transistor 30 is connected through the voltage supply 26 to the output of the class-G amplifier, while the base of transistor 36 is connected through a voltage supply 28 to the output of the class-G amplifier. Transistor 32 has a base connected to the output of pull-up current mirror 18, while the base of transistor 34 is connected to the output of pull-down current mirror 20. A first low voltage power supply Vsp1 is connected through diode 38 to the collector of transistor 32 at node p, while a second low voltage power supply Vsm1 is through diode 40 to the collector of transistor 34 at node m. Capacitors 22 (CBCp) and 24 (CBCm) represent the parasitic capacitance that loads the terminals of diodes 14 and 16. A feedback resistor 42 is typically connected from the output of the class-G amplifier to the inverting input (xe2x88x92input).
The amplifier of FIG. 1 operates as a class-G amplifier in that the collector voltages of transistors 32 and 34 are provided by one of transistor 30 and diode 38, or one of transistor 36 and diode 40, respectively. That is for small output voltages the diodes 38 and 40 are forward biased and load current flows through diode 38 or 40, and transistors 30 and 36 are biased off. When the output voltage exceeds Vsp1+VBExe2x88x92VDxe2x88x92VBp for positive swing (where VBE is the base to emitter voltage of transistor 30, VD is the diode voltage for diode 38, and VBp is the voltage supply 26 voltage) or xe2x88x92Vsm1+VBE+VD+VBm for negative swings (where VBE is the base to emitter voltage of transistor 34, VD is the diode voltage for diode 40, and VBm is the voltage supply 28 voltage), transistor 30 or 36 will turn on and divert output current from one of the low voltage supplies VspL or Vsm1 toward one of the high voltage supplies Vsph or Vsmh. Thus, small signals at the output will cause current to be drawn from the low supplies. In the case of Digital Subscriber Line (DSL) waveforms, only 1-3% of signal swings will draw power from the high supplies, and overall power consumption is minimized.
At low output levels, the dominant compensation pole for the amplifier of FIG. 1 is at 1/(2 RF*(CBCP+CBCm)), where RF is the value of feedback resistor 42. The voltage at the collectors of transistors 32 and 34 does not move much with small outputs, so capacitors 22 (CBCp) and 24 (CBCm) load the xe2x80x9cgain nodexe2x80x9d between the terminals of diodes 14 and 16. When large outputs occur, however, transistors 30 and 36 drive the collectors of transistors 32 and 34. For large positive outputs, for instance, transistor 30 provides a voltage at the collector of transistor 32 that follows the output. Thus, capacitor 22 is now driven with similarly changing voltages at both terminals and draws almost no AC current. Capacitor 22 therefore no longer adds its capacitance as compensation in the above equation. The fed-back pole is moved upward in frequency. A similar operation occurs with large negative outputs. With a large negative output, transistor 36 provides a voltage at the collector of transistor 34 that follows the output. Thus, capacitor 24 is now driven with similarly changing voltages at both terminals and draws almost no AC current. Capacitor 24 therefore no longer adds its capacitance as compensation.
Unfortunately, at the higher pole frequency additional phase lag exists in the current mirrors and all other transistors, and the circuit is more likely to oscillate. Additionally, a sudden change in frequency response occurs in this supply crossover region, causing a transient response with each traversal of the region. This leads to higher output distortion.
In accordance with the present invention, referring to FIG. 2, an improved Class-G amplifier is provided by adding a first capacitor 82 between the input of current mirror 18 and node p, and by adding a second capacitor 84 between the input of current mirror 20 and node m. The added capacitors 82 and 84 can be sized to stabilize frequency responses when high power supplies are enabled. The added capacitors 82 and 84 further function to reduce transient currents during switching through the crossover points between upper and lower power supplies.